Self-calibrating precision timing circuit and method for a laser range finder

ABSTRACT

A highly precise range measurement instrument is made possible through the use of a novel and efficient precision timing circuit which makes use of the instrument&#39;s internal central processing unit crystal oscillator. A multi-point calibration function includes the determination of a “zero” value and a “cal” value through the addition of a known calibrated pulse width thereby providing the origin and scale for determining distance with the constant linear discharge of capacitor.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application claims priority from and is a continuation ofU.S. patent application Ser. No. 09/234,724 filed Jan. 21, 1999, (nowU.S. Pat. No. 6,057,910), a continuation of U.S. patent application Ser.No. 08/918,396 filed Aug. 26, 1997 (now U.S. Pat. No. 5,880,821), acontinuation of U.S. patent application Ser. No. 08/717,635 filed Sep.23, 1996 (now U.S. Pat. No. 5,703,678), a continuation of U.S. patentapplication Ser. No. 08/375,941 filed Jan. 19, 1995 (now U.S. Pat. No.5,574,552), each incorporated herein by reference, from which priorityunder 35 U.S.C. §120 is claimed and which are assigned to the assigneeof the present application.

The present invention is related to those disclosed and claimed in U.S.patent applications Ser. No.: U.S. Pat. No. 5,652,651 for: “Laser RangeFinder Having Selectable Target Acquisition Characteristics and RangeMeasuring Precision”; and [Attorney Docket No. 35014.8308] for:“Automatic Noise Threshold Determining Circuit and Method for a LaserRange Finder”, all filed concurrently with U.S. patent application Ser.No. 08/375,941, from which priority under 35 U.S.C. §120 is claimedherein, all of which are assigned to the assignee of the presentinvention, Laser Technology, Inc., Englewood, Colo., the disclosures ofwhich are hereby specifically incorporated by this reference.

BACKGROUND OF THE INVENTION

The present invention relates, in general, to the field of distance orrange measuring equipment. More particularly, the present inventionrelates to a laser based range finder which may be inexpensivelyproduced yet provides highly accurate precision range measurements of upto 1000 yards or more with a resolution of less than 1 yard. The laserrange finder herein disclosed has a number of user selectable targetacquisition and enhanced precision measurement modes which may be viewedon an in-sight display during aiming and operation of the instrument.Extremely efficient self-calibrating precision timing and automaticnoise threshold circuits incorporated in the design provide a compact,low-cost, highly accurate and reliable ranging instrument for amultitude of uses.

Laser based distance and range measuring equipment have been used for anumber of years to provide extremely accurate distance measurements to aremote target or object. A representative instrument is the Criterion™100 laser range finder developed and marketed by Laser Technology, Inc.,assignee of the present invention. Although a highly accurate andreliable device, its great distance ranging capability and inherentcomplexity translates to a cost and form factor most suitable only forcertain specific applications. A need therefore exists for a laser basedrange finder of perhaps more limited range, which can be economicallymanufactured as a rugged, compact unit to provide accurate distancemeasurement capabilities in other less stringent types of applications.

SUMMARY OF THE INVENTION

Herein disclosed is a precise, yet accurate and reliable laser rangefinder which may be economically produced and is adapted to individualportable use in a unit potentially weighting less than a pound with anon-board battery based power supply. Moreover, the compact instrumentherein provided has a number of user selectable target acquisitionoperational modes which may be invoked depending on the distance, typeand reflectivity of the target being sighted.

Though the use of an in-sight display, distance or range information canbe shown while the user may also view and select the instrument's modeof operation through successive actuations or a push button mode switchwhile simultaneously sighting the target object. A precision mode ofoperation may also be invoked in which an even more precise measurementto an object may be achieved following an initial measurement togetherwith the visual indication of a “precision flag” on the in-sightdisplay.

A highly precise range measurement is made possible through the use of anovel and efficient timing circuit which makes use of the instrument'sinternal central processing unit crystal oscillator. A likewise uniqueautomatic noise threshold determining circuit allows for instrumentoperation with a low signal-to-noise ratio to optimize sensitivity andperformance in conjunction with a processor based pulse discriminationprocedure which, nevertheless assures accurate range measurements.

The unit herein disclosed can be utilized in a multitude of endeavorsincluding such recreational activities as golf where it can be utilizedto very accurately determine the distance to a flag or pin as well as totrees and other natural objects. The principles of the invention arefurther applicable to the design or a laser based “tape measure” whereranges can be precisely measured with resolutions of on the order of aninch or less.

Specifically disclosed herein is a self-calibrating, precision timingcircuit and method for determining a range to a target based upon aflight time of a pulse toward the target. The circuit comprises meansfor initially establishing first and second reference voltage levelstogether with means for unclamping the second reference voltage leveland means for allowing the second reference voltage level to thendiminish at a first rate to the first reference voltage level. Furtherprovided are means for storing a first reference time extending from thestep or unclamping until the first and second reference voltage levelsare determined to be equal. Means are also provided for thenre-establishing the first and second reference voltage levels togetherwith means for again unclamping the second reference voltage level.Additional means are provided for increasing the second referencevoltage level at a second higher rate than the first rate for apredetermined period of time to establish a third reference voltagelevel together with means for then allowing the third reference voltagelevel to diminish at the first rate to the first reference voltage levelat which time, a second reference time extending from the step of againunclamping until the first and third reference voltage levels are equalis additionally stored. The first and second reference voltage levelsare again re-established and the second reference voltage level isfurther unclamped. Means are provided for again increasing the secondreference voltage level at the second higher rate for a period of timerelated to the flight time of the pulse to the target to establish afourth reference voltage level, together with means for then allowingthe fourth reference voltage level to diminish at the first rate to thefirst reference voltage level. A third reference time extending from theunclamping of the second reference voltage level until the first andfourth reference voltage levels are equal is then stored and the rangeto the target may be computed as proportional to the quantity of the(third reference time minus the first reference time) divided by thequantity of the (second reference time minus the first reference time).

In a particular embodiment the establishing means may comprise atransistor switch for coupling a capacitor to a source of the secondvoltage while the unclamping means may comprise a second transistorswitch for decoupling the capacitor from the second voltage source. Theallowing means may comprise a third transistor switch coupling aresistor to the capacitor to bleed off the charge therefrom.

The means for increasing the second reference voltage level may comprisemeans for applying a charge to the capacitor at the second rate and thepredetermined time period specified may be determined by reference to acrystal oscillator. In a particular embodiment, the second charging ratemay be substantially 1000 times the first discharging rate.

DETAILED DESCRIPTION OF THE DRAWINGS

The foregoing and other features and objects of the present inventionand the manner of attaining them will become more apparent and theinvention itself will be best understood by reference to the followingdescription of a preferred embodiment taken in conjunction with theaccompanying drawings, wherein:

FIG. 1 is a simplified logic block diagram of a laser range finder inaccordance with the present invention illustrating the significantfunctional aspects thereof, inclusive of a laser signal transmitting andreceiving section, central processing unit and the precision timing andautomatic noise threshold sections thereof;

FIG. 2 is a detailed schematic diagram of the laser transmit section ofFIG. 1 illustrating, inter alia, the laser signal producing diode andthe associated driving and reference signal producing circuitry;

FIG. 3 is an additional detailed schematic diagram of the laser receivesection of FIG. 1 illustrating, inter alia, the laser signal receivingdiode, transimpendance amplifier and the precision comparator forestablishing the V_(threshold) and RX(Out+) signals for the precisiontiming and automatic noise threshold circuits;

FIGS. 4 and 5 are further detailed schematic diagrams of the precisiontiming section of the laser range finder of FIG. 1 illustrating thecircuit nodes for establishing the voltages V₁ and V₂ during the zero,calibration (“CAL”) and laser firing phases of operation;

FIG. 6 is an additional detailed schematic diagram of the centralprocessing unit (“CPU”) portion of the laser range finder of FIG. 1illustrating the CPU, associated oscillator and the in-sight liquidcrystal display (“LCD”) for displaying measured distances to an operatorof the laser range finder in addition to the various signals foroperative association with the precision timing and automatic noisethreshold sections thereof;

FIGS. 7A, 7B and 7C are individual graphic representations of thevoltages V₁ and V₂ of certain of the precision timing section circuitnodes during the zero, calibration and laser firing phases of operationfrom which the values Zero_(TIME), Cal_(TIME) and Laser_(TIME) arederived to enable rapid and accurate calculation of the distance to anobject from the laser range finder; and

FIG. 8 is a final detailed schematic diagram of the automatic noisethreshold section of the laser range finder of FIG. 1 illustrating thevarious components thereof as well as the signals coupling the same tothe laser receive section and CPU.

DESCRIPTION OF A PREFERRED EMBODIMENT

With reference now to FIG. 1, a logic block diagram of a laser rangefinder 10 in accordance with the present invention is shown. The laserrange finder 10 includes, in pertinent part, a main power supply unit(“PSU”) 12 as operatively controlled by a trigger switch 14. The mainpower supply unit 12 is coupled to a high voltage (“HV”) power supplyunit 16 for supplying operating power in conjunction with the main powersupply unit 12 to a laser transmit section 18.

The laser transmit section 18 activates a laser emitting diode 20 fordirecting a laser signal toward an object in the operation of the laserrange finder 10. The laser transmit section 18 also supplies a /FIREsignal to the central processing unit (“CPU”) section 28 as will be morefully described hereinafter.

The main power supply unit 12 also supplies operating power to a laserreceive section 22 which further has an input a signal generated by alaser receiving diode 24 as the laser signal emitted from the laseremitting diode 20 is reflected from an object back thereto. The laserreceive section 22 supplies a V_(threshold) signal and RX(OUT+) signalto an automatic noise threshold section 36 and a precision timingsection 34 both of which will be described in more detail hereinafter.

The CPU section 28 receives as one input a signal from a mode switch 26by means of which an operator can change the operating mode andfunctional operation of the laser range finder 10. An oscillator 30supplies a clocking signal to the CPU section 28 as well as to theprecision timing section 34. The CPU section 28 provides an outputindicative of the distance from the laser range finder 10 to an objectas sighted through a viewing scope thereof on an in-sight liquid crystaldisplay (“LCD”) 32.

The precision timing section 34 provides a number of signals to the CPUsection 28 including a TIMER and /RX DETECT signals as shown andreceives a {overscore (RUN)}/CLAMP signal back therefrom. The CPUsection 28 provides a number of signals to the precision timing section34 including a HOLD OFF, NORM/CAL, /RESET, and a CAL DITHER signal. Theautomatic noise threshold section 36 also receives a number of inputsfrom the CPU section 28 including a number of noise set (“NSET”) signalsand a REFLECTION MODE signal to operatively control its function.

With reference additionally now to FIG. 2, the laser transmit section 18is shown in more detail. The laser transmit section 18 receives atransmit (“TX”) BIAS signal on supply line 50 of approximately 110 to140 volts for application through resistor 52 to the emitter oftransistor 54. The emitter of transistor 54 is coupled to its base bymeans of a resistor 58 which also couples the collector of transistor 56to resistor 52. The emitter of transistor 56 is connected to circuitground on ground line 60. A capacitor 62 couples the emitter oftransistor 54 to the cathode of the laser emitting diode 20 which hasits anode also connected to circuit ground 60. An additional diode 64 iscoupled in parallel with the laser emitting diode 20 having its anodeconnected to the cathode of the laser emitting diode 20 and its cathodeconnected to circuit ground 60. A resistor 66 is placed in parallel withthe laser emitting diode 20 and the diode 64.

A source of +5 volts is also received by the laser transmit section 18on supply line 68 through resistor 70. Resistor 70 is coupled to theemitter of transistor 72 as well as to circuit ground 60 through acapacitor 74. A resistor 76 couples the emitter of transistor 72 to itsbase which is coupled through resistor 78 to line 80 for supplying a/FIRE signal to the CPU section 28 (shown in FIG. 1).

An additional diode 82 has its anode connected to the collector oftransistor 72 and its cathode coupled to circuit ground 60 throughresistor 86. A capacitor 84 couples the cathode of diode 82 to thecommon connected collector of transistor 54 and base of transistor 56.The common connected collector of transistor 54 and base of transistor56 is coupled through a voltage divider network comprising resistor 88and resistor 90 to circuit ground. A resistor 92 coupled betweenresistor 88 and resistor 90 provides a REF signal on line 94 forapplication to the precision timing section 34 (shown in FIG. 1).

With reference additionally now to FIG. 3, the laser receive section 22is shown in more detail. The output signals of the laser receive section22 are the signals RX(OUT+) and V_(threshold) provided on lines 100FIGS.4, 8; and 102FIG. 8 respectively for application to the precision timingsection 34 and automatic noise threshold section 36 as previously shownin FIG. 1. A source of +50 volts providing a receive (“RX”) BIAS signalis input to the laser receive section 22 from the HV power supply unit16 on supply line 104. A low pass filter network 106 comprisingresistors 108 and 112 in conjunction with capacitors 110 and 114 couplesthe supply line 104 to circuit ground 60 to provide a bias signal to thecathode of the laser receiving diode 24. The laser receiving diode 24has its anode connected to the base of transistor 118 which, inconjunction with transistor 120, 122, and 124 comprises a transimpedanceamplifier 116 providing an output on node 126 which is capacitivelycoupled to the “+” input of a precision comparator 134. A source of +5volts is input to the laser receive section 22 from the main powersupply unit 12 (shown in FIG. 1) for input to the transimpedanceamplifier 116 through a low pass filter comprising resistor 130 andcapacitor 132. The +5 volt RX supply voltage is also coupled to the v+input of the precision comparator 134 through resistor 136 and iscoupled to circuit ground through capacitor 138. The “+” input of theprecision comparator 134 is connected between the plus 5 volt RX voltagesource and circuit ground 60 through the node intermediate resistor 142and resistor 144.

The precision comparator 134 which may, in a preferred embodiment,comprise a MAX 913 low power precision transistor-transistor logic(“TTL”) comparator available from Maxim Integrated Products, Inc.,Sunnyvale, Calif., has its “v−”, “LE” and ground (“GND”) inputsconnected to circuit ground 60 as shown. A capacitor 146 couples the “−”output of the precision comparator 134 to circuit ground 60 as shown.The “O+” output of the precision comparator 134 is supplied through aresistor 148 to line 100 to provide the RX(OUT+) signal while the “−”output of the precision comparator 134 is supplied through resistor 150to line 102 to provide the V_(threshold) signal.

With reference additionally now to FIG. 4, a portion of the precisiontiming section 34 (shown in FIG. 1) is illustrated. A CPU clock (“CLK”)signal is input to the precision timing section 34 on line 152 to theCLK input of a serial in/parallel out shift register 160 from theoscillator 30 as previously shown in FIG. 1. An additional input to theshift register 160 is received on line 154 comprising a NORM/CAL signalfrom the CPU section 28 to the data set B (“DSB”) input thereof. Theactive low clear (“{overscore (CLR)}”) input and DSA input are held highas shown.

An additional input to the precision timing section 34 is received fromthe CPU section 28 (shown in FIG. 1) on line 156 comprising a /RESETsignal for input to the reset (“{overscore (R)}”) inputs of D typeflip-flop 158 and flip-flop 162. The {overscore (Q)} output of flip-flop158 is supplied as one input to an invertor comprising a portion of aNAND Schmitt trigger 168 through a low pass filter comprising resistor164 and capacitor 166 as shown. The remaining input to the invertor 168is connected to a source of +5 volts.

A resistor 172 couples a source of +5 volts to the collector oftransistor 174 having its emitter coupled to circuit ground. Thecollector terminal of transistor 174 is coupled through capacitor 170 tothe input of the invertor 168 coupled to the {overscore (Q)} output offlip-flop 158. Transistor 174 has its based coupled to circuit groundthrough resistor 176 and receives a HOLD OFF signal on node 178 receivedfrom the CPU section 28.

The flip-flop 158 receives an input to its CLK terminal on line 94comprising the REF output signal from the laser transmit section 18(shown in FIG. 1). Its data (“D”) input is coupled to a source of +5volts and the Q1 output of the shift register 160 is provided to theactive low set (“{overscore (S)}”) input as shown. The Q output offlip-flop 158 is supplied as one input to a transmit gate 204 having itsother input coupled to the output of an invertor comprising anadditional NAND Schmitt trigger 202. Invertor 202 has one inputconnected to a source of +5 volts and another input connected to the Qoutput of flip-flop 162. Flip-flop 162 has its {overscore (S)} inputcoupled to the Q7 output of shift register 160 and its D input connectedto the output of invertor 168. The {overscore (Q)} output of flip-flop162 is supplied on line 184 to comprise a /RX DETECT signal for input tothe CPU section 28 (shown in FIG. 1). The flip-flop 162 has its CLKinput connected to line 100 for receiving the RX(OUT+) signal from thelaser receive section 22 (shown in FIG. 1) which is also supplied as oneinput to NAND Schmitt trigger 180. The other input of NAND Schmitttrigger 180 is connected to line 184 through resistor 182 and coupled tocircuit ground through capacitor 186. The output of Schmitt trigger 180is supplied to the base electrode of transistor 200 which has itscollector terminal coupled to circuit ground. Line 196, comprising ananalog-to-digital (“A/D”) POWER CORRECTION signal is supplied to theemitter terminal of transistor 200 through resistor 198 as well as tothe collector terminal of transistor 190 which is coupled to circuitground through capacitor 194. The /RESET signal on line 156 is suppliedto the base terminal of transistor 190 through resistor 188. A source of+5 volts is connected to the emitter of transistor 190 as well asthrough resistor 192 to the base of transistor 190 to provide anoperating bias.

Referring additionally now to FIG. 5, the remaining portion of theprecision timing section 34 (shown in block form in FIG. 1) isillustrated. The HOLD OFF signal output from CPU section 28 to theprecision timing section 34 is supplied on line 258 through resistor 256to node 178 for input to the base of transistor 174 (shown in FIG. 4).

The output of transmit gate 204 appearing on node 206 is suppliedthrough resistor 208 to the base terminal of transistor 210. A source of+5 volts is supplied to the emitter terminal of transistor 210 throughthe series connection of resistor 216 and resistor 222. The nodeintermediate resistors 216 and 222 is coupled to circuit ground throughthe parallel combination of capacitors 218 and 222 as well as to theoutput of comparator 236 through resistor 246 to provide a TIMER signalon line 250 for input to the CPU section 28 as will be more fullydescribed hereinafter. The source of +5 volts is also connected to thebase terminal of transistor 210 through the series connection ofresistors 216 and 224. A V₁ node 228 at the common connected base oftransistor 212 and emitter of transistor 214 is coupled through a sourceof +5 volts through resistor 216 and resistor 226. Node 228 is connectedthrough resistor 230 to V₂ node 232 which, in turn, is connected tocircuit ground through resistor 240. A capacitor 238 couples V₁ node 228to circuit ground. V₂ node 232 is connected to the “−” input ofcomparator 236. V₁ node 228 is connected to line 254 from the CPUsection 28 (shown in FIG. 1) to receive the CAL DITHER signal throughresistor 252.

The collector terminal of transistor 210 is coupled to the collectorterminals of transistors 212 and 214 as well as to the “+” terminal ofcomparator 236 which, in turn, is coupled to circuit ground throughcapacitor 244. A {overscore (RUN)}/CLAMP signal output from the CPUsection 28 (shown in FIG. 1) is furnished on line 260 through resistor248 for input to the base terminal of transistor 214.

With reference additionally now to FIG. 6, the CPU section 28 is shownin greater detail. The CPU section 28 comprises, in pertinent part, amicrocomputer 270 which may, in a preferred embodiment, comprise aST6240 device. An 8 megaHertz (“MHz”) crystal 274 forms a portion of theoscillator 30 for providing an oscillator (“OSCIN”) and oscillator out(“OSCOUT”) signal to the microcomputer 270 as well as supplying a CPUCLK signal on line 152 for input to the precision timing section 34 aspreviously described. The VDD input of microcomputer 270 is coupled to asource of +5 volts and the /RESET input thereof is held high throughpull up resistor 276 which is coupled to circuit ground throughcapacitor 278. Output from the microcomputer 270 is taken on a displaybus 280 comprising the communication (“COM”) lines COM 1-COM 4 andS16-S28 lines for input to the LCD display 32.

An A/D LOW BATTERY signal, a TRIGGER signal, and a POWER CONTROL signalare input to the microcomputer 270 on lines 284, 286, and 288respectively. The A/D LOW BATTERY signal on line 284 is also supplied tothe “−” input of comparator 296 which is coupled to circuit groundthrough capacitor 304. The “+” input of comparator 296 is coupled to asource of +5 volts through resistor 298 which is also coupled to circuitground through the parallel combination of resistor 300 and capacitor302. The output of comparator 296 appearing on line 306 provides aSHUTDOWN signal for the laser range finder 10 in the event the onboardbattery voltage drops below a predetermined limit.

The microcomputer 270 supplies the HOLD OFF signal on line 258, the{overscore (RUN)}/CLAMP signal on line 260, the CAL DITHER signal online 254, the /RESET signal on line 156 and the NORM/CAL signal on line154 for input to the precision timing section 34 as has been previouslydescribed. The microcomputer 270 receives as outputs from the precisiontiming section 34 the /RX DETECT signal on line 184 and the TIMER signalon line 250. Additional inputs to the microcomputer 270 are the /FIREsignal on line 80 from the laser transmit section 18 (shown in FIG. 1)as well as the A/D POWER CORRECTION signal on line 196 from theprecision timing section 34 (as shown in FIG. 4). A MODE input signal online 294 is received from the mode switch 126 which is otherwise held toa +5 volts through resistor 292. Microcomputer 270 supplies an NSET1 andNSET2 signal on lines 308 and 310 respectively as well a REFLECTION MODEsignal on line 312 for input to the automatic noise threshold section 36(as shown in FIG. 1).

In overall operation, a reference signal (REF) on line 94 is generatedby the laser transmit section 18 (shown in FIG. 2) when the laser rangefinder 10 is fired by placing a current pulse through the laser emittingdiode 20 in response to manual actuation of the trigger switch 14. TheREF signal on line 94 is derived from the current placed through thelaser emitting diode 20 and not from the light pulse itself and issufficiently precise for accurately indicating the time of the laserfiring. The REF signal is ultimately input to the CLK input terminal offlip-flop 158, which has its Q output coupled to the transmit gate 204,which then turns on the current switch comprising transistor 210, andstarts charging the capacitor 244. When the receive pulse (RX(OUT+) online 100 comes back from the laser receive section 22 (shown in FIG. 3),it triggers the flip-flop 162 at its CLK input. Flip-flop 162 has its Qoutput coupled to the input of invertor 202 which then shuts thetransmit gate 204 off, stopping the current pulse. At this point, aconstant current sink discharges capacitor 244. In this manner,capacitor 244 is charged up with a relatively large current (on theorder of 10 milliamps), and later discharged with a small current (onthe order of 10 microamps) applied over the entire flight time of thelaser pulse from its firing from the laser emitting diode 20 to itsreflection from a target back to the laser receiving diode 24. Becausethe laser range finder 10 is intended for a shorter maximum range thanother laser based range finding instruments, the use of this techniquedoses not require a separate counting oscillator followed by aninterpolation operation and the entire flight time is essentiallystretched by a factor of 1000 and then the stretched result is counted.By charging capacitor 244 at a fast rate and then discharging it andthen monitoring the time it takes to discharge, the flight time isexpanded so that the slower clock in the CPU section 28 can then countit accurately. The microcomputer 270 utilized in the CPU section 28 hasa 1.5 microsecond resolution and, because the incoming flight time hasbeen expanded by a factor of 1,000 on the input side to the precisiontiming section 34, it is the equivalent of a 1.5 nanosecond resolution,which corresponds to a measurement resolution for the laser range finder10 of on the order of nine inches. Therefore, given that the laser rangefinder 10 is intended to be a one-yard instrument with a nine-inchresolution, sufficient resolution is provided to be able to measuredistances up to a thousand yards to a one-yard accuracy.

The precision timing section 34 of the laser range finder 10 has threedistinct modes of operation including a zero calibration, fixed pulsewidth calibration and laser measurement function as will be more fullydescribed hereinafter. The portion of the precision timing section 34comprising transistors 210, 214, and 212 (shown in FIG. 5) is theessence of the integrating flight time expander. Transistor 210functions as a current switch which is turned on for the duration of thelaser flight time in the laser mode of operation and is also turned onfor the duration of whatever calibration pulse is placed into it duringthe calibrate mode. In the latter instance, a calibration pulse issupplied by the shift register 160 via flip-flop 158 and the start andend of the calibration pulse is gated via transmit gate 204 to actuallyturn the transistor 210 on and off in order to function as a currentsource, typically sourcing 10 milliamps of current. It should be notedthat prior to turning transistor 210 on, transistor 214 must first beturned off and, when the system is in the reset state ready to start thewhole measurement sequence, transistor 210 is off. Transistor 212, whichis the current sink in the system, is always on, and typically sinks onthe order of 10 microamps of current. In the reset condition, transistor214 is on, and that clamps the voltage at the top plate of capacitor 244to a voltage level designated a V1 at node 228. A voltage V2 is definedas the voltage at node 232 at the “−” input of comparator 236. It shouldalso be noted that a metal oxide semiconductor field effect transistor(“MOSFET”) may be utilized for transistor 40 and would exhibit a muchlower offset than the bipolar device shown. However, due to the lowercost of bipolar transistors and the fact that any offset cancels duringthe processing of the signal, a bipolar transistor is entirely adequatefor this purpose.

When transistor 214 is on, the voltage on the positive plate ofcapacitor 244 is clamped to voltage V1, plus a fixed offset due to thetransistor 210, which is small and typically on the order of 50millivolts. During the zero calibration function, transistor 214 isturned on by holding the {overscore (RUN)}/CLAMP signal on line 260high, thereby applying a positive current to its base through resistor248. To initiate the zero calibration, the TIMER signal on line 250 isasserted and supplied to the microcomputer 270 of the CPU section 28.Utilizing the ST6240 unit shown in FIG. 6, when the microcomputer TIMERpin is held high, the device is counting. Conversely, the microcomputerstops counting when the pin is allowed to go low. In operation, theoutput comparator 236, determines whether or not the voltage at the topplate of capacitor 244 is greater or less than V2, and its outputdetermines whether the TIMER pin on the microcomputer 270 is high orlow. In the normal reset condition, the output of the comparator 236 ishigh, which means the time is active. In sequence, the microcomputer 270initiates the TIMER function and then turns off transistor 214 bylowering the control signal {overscore (RUN)}/CLAMP on line 260, tounclamp capacitor 244. Capacitor 244 then starts discharging towardszero due to the current being drained out of it via transistor 212 at arate of about ten microamps. When it has discharge such that the chargeremoved drops the voltage V1 at node 228 to the level of V2, the outputof the comparator 236 changes state to stop the TIMER function. (In theparticular embodiment shown, V1 is typically on the order of 1.0 voltsand V2 is about 0.9 volts.) The microcomputer 270 of the CPU section 28now has a count value that relates to the amount of time it takes forcapacitor 244 to discharge from V1 down to V2. This process is repeatedseveral times and the result is averaged. Typically ten iterations maybe performed with the results accumulated and an average time computed.

As shown particularly with respect to FIG. 5, the CAL DITHER signal online 254 is applied to the base terminal of transistor 212 and isutilized during both the zero calibration and fixed pulse widthcalibration times and incorporates a relatively high value resistor 252.The CAL DITHER signal allows for the introduction of a deliberatelycontrolled change in the discharge current in order that the resultantcount will vary slightly such that when the total counts are averagedtogether, a finer resolution is produced than would be the case merelyusing a fixed current to get the same count value. An adjustment of onepart in about a thousand is provided during the zero calibration andfixed pulse width calibration modes because the finite resolution of themicrocomputer 270 timer otherwise provides discreet timing intervals of1.5 nanoseconds which would only provide distance measurement resolutionof approximately one yard. In operation, the zero calibration count inthe microcomputer 270 will typically be about 150 while in the fixedpulse width calibration mode it will be on the order of 900. The flighttime count during the laser mode of operation can be anything from closeto the zero calibration value to about 4500.

For example, during the zero calibration mode, the count value in themicrocomputer 270 might be 150 but there is no way of knowing just howclose the count actually is to 149 to 151. By utilizing the CAL DITHERsignal to force the count over a couple of count boundaries (forexample: 150, 150, 150, 151, 151, 152) the resolution of the counter maybe effectively raised by a factor of two without having to utilizeadditional fine counters. In the embodiment shown, the resultantresolution is sufficient to maintain calibration to plus or minus oneyard over a range of one thousand yards or less. Althoughimplementations may vary, the CAL DITHER signal may be held high forfive out of ten pulses and low for the remainder to provide theforegoing resolution enhancement.

Due to the fact that the actual laser flight time varies due to noise inthe laser pulses and variability in target aiming, there is generallyenough scatter in the measured laser flight time such that it coversmore than one clock boundary and so will automatically average to ahigher resolution through the use of the precision timing section 34without invoking the CAL/DITHER function in the laser mode of operation.

With reference additionally now to FIGS. 7A, 7B and 7C, the operation ofthe precision timing section 34 is shown in the zero calibration, fixedpulse width calibration and laser measurement function modes ofoperation respectively. In its normal state, the voltage on the topplate of capacitor 244 is clamped at V1, and at a time T₀, the precisiontiming section 34 will initiate the TIMER by changing the output stateof comparator 236 to the logic high state. After a very short fixednumber of instructions later shown as T₁, the clamp transistor 214 willbe turned off and the voltage on capacitor 244 will begin dischargingslowly until that voltage crosses V2 at time T₃ when the output ofcomparator 236 will change state. In essence, during the zerocalibration process, transistor 210 is never turned on therebydetermining the timing conditions of what would effectively be a zeroflight time. Therefore, if there is no charge current applied tocapacitor 244, T₃−T₀ zero is the time that would be in the microcomputer270 and the timer in whatever units they operate, which is usuallydependent on the CPU section 28 crystal frequency. In the embodimentshown, the microcomputer 270 utilizes an 8 MHz crystal and the internaltimer has a 1.5 microsecond resolution resulting in a count of about150.

During the fixed pulse width calibration process (shown particularly inFIG. 7D) at time T₄, once again the microcomputer 270 stops the TIMERand a short time later at T₅ it releases the clamp. At T₆, a known pulsewidth is applied to the base terminal of transistor 210 which isprecisely derived from the main oscillator 30 as applied to the CLKinput of the shift register 160. The signal applied to the CLK input ofthe shift register 160 directly tracks the main oscillator 30 and theserial data input to the shift register 160 is a logic line 154 from theCPU section 28 designated NORM/CAL. When the NORM/CAL signal is high,the precision timing section 34 is in its normal mode of operation and,when it drops to a logic low state, the fixed pulse width calibrationfunction is initiated. Thereafter, typically about fifty microsecondslater, at time T₆ the NORM/CAL signal on line 154 will be dropped low.It should be noted that during both the zero and the fixed pulse widthcalibration modes, the logic reset signal /RESET on line 156 is heldlow, its active state. In the logic low state the two flip-flops 158,162 determine whether the input signal comes from shift register 160which generates the fixed pulse width or whether it comes from the REFand RX(OUT+) signals an relates to an actual laser flight time. The/RESET signal is generally held low at all times during the fixed pulsewidth calibration process so that any noise on the RX(OUT+) receive line100 will not accidently clock flip-flop 162 and therefore trigger theprecision timing section 34 resulting in an indeterminate time periodmeasurement invalidating the calibration. The reset state for the Qoutputs of flip-flops 158, 162 is low but is high for the {overscore(Q)} outputs. Therefore, the {overscore (Q)} outputs can not be directlydriven with the reset circuit and must be driven off the Q outputs inboth cases which introduces a small fixed offset delay which must beaccounted for later. As soon as the NORM/CAL signal on line 154 isdropped low, which occurs approximately 50 microseconds after the clamphas been released, the low signal propagates through the shift register160 precisely with the main oscillator 30 clock. The Q0 output of theshift register 160 is the first to be triggered but is not used becauseit is used to synchronize with the incoming signal. The Q1 is then thefirst output of the shift register 160 to be utilized and on everypositive edge of the clock the zero signal that is applied into theserial input will propagate one state of the shift register 160 from Qzero to Q7. Therefore, the Q1 output will go low first, and as soon asthat output goes low, the set line input {overscore (S)} forces the Qoutput of flip-flop 158 to go high since the Q output of flip-flop 162is in the low state. As a result, logic level ones appear at the twoinputs of the transmit gate 204, which turns on the current switchtransistor 210. Exactly six clocks later, the same thing happens withflip-flop 162 which has its {overscore (S)} input coupled to the Q7output of the shift register 160. As the Q output of flip-flop 162 goeshigh, the output of the invertor 202 goes low, and the transmit gate 204will be turned off. At this point the count pulse will stop meaning thatthe fixed width pulse feeding the current switching circuit at theoutput of the transmit gate 204 is precisely six clock cycles. The timedifference between the Q1 and Q7 outputs of the shift register 160 isexactly 750 nanoseconds when utilizing an 8 MHz oscillator 30 applied toits CLK input. The invertor 202 adds an additional delay of about 10nanoseconds for a total of delay of about 760 nanoseconds which variesonly slightly with temperature, perhaps one or two nanoseconds, yetstill provides sufficient precision for measurements of less than oneyard resolution.

Transistor 210 is then turned on for a period of time between T₆ and T₇to enable the capacitor 244 to charge very rapidly and then discharge atthe same rate as has been previously shown with respect to FIG. 7A. AsV1 reaches the level of V2 the TIMER signal goes low at Time T₈. Thefifty microsecond delay between the unclamping at T₅ and T₆ is to allowthe clamp transistor 214 to turn off fully since it is a relativelyinexpensive bipolar device. If a MOSFET were used instead, its turn offwould be virtually instantaneous and the additional delay it introducedwould not be a problem because the microcomputer 270 couldn't issue thenext instruction quickly enough. Utilizing a bipolar device,approximately 20 microseconds are required for the discharge to becomelinear and the slope of the discharge curve between T₇ and T₈ is thenidentical to the slope from T₁ to T₃ in the zero calibration mode exceptfor the step due to the charging of capacitor 244. As a consequence, thevalue of ZERO_(TIME) equals T₃ minus T₀ and the value of CAL_(TIME)value equals the time due to the CAL_(TIME) value not due to theZERO_(TIME) value, which is, T₈ minus T₄ minus the ZERO_(TIME) value or,T₈ minus T₃.

In essence then, very small flight times are effectively disregarded andthe value of CAL_(TIME) is known. Therefore, with the zero calibrationfunction and the addition of a known calibrated pulse width, the timedelay at zero is known together with the time delay for the known pulsewidth providing the origin and scale for determining distance with aconstant linear discharge of capacitor 244.

With particular reference additionally to FIG. 7c, the operation of theprecision timing section 34 is shown in the laser measurement mode ofoperation. The laser measurement operation is essentially the same asthe fixed pulse width calibration mode except that the NORMAL/CAL signalon line 154 to the shift register 160 is held high and the /RESET signalon line 156 is taken high at time T₉ to enable the flip-flops 158, 162to trigger. At time T₁₀ the timer is started and at T₁₁, (at preciselythe same relationship T₁₁ minus T₁₀ equals T₅ minus T₄ equals T₁ minusT₀) the clamp is released. There is normally a fifty microsecond waitand then the laser pulse is fired when the microcomputer 270 asserts the/FIRE signal on line 80 to initiate the firing sequence. Upon firing thelaser emitting diode 20, the laser transmit section sends the REF signalon line 94 to the CLK input of flip-flop 158 of the precision timingsection 34. This opens the transmit gate 204 which turns on the currentsource transistor 210, which, in turn, charges capacitor 244 at a knownrate.

When the reflected laser pulse is detected by the laser receiving diode24 of the laser receive section 22 (shown in FIG. 3), the RX(OUT+)signal on line 100 is directed to the CLK input of flip-flop 162. The Qoutput signal of flip-flop 162 is inverted by invertor 202 which turnsoff the transmission gate 204 so that the current source transistor 210is on for the flight time duration of the laser pulse to chargecapacitor 244 to a level determined by the timer during that flighttime. The charge applied to the capacitor 244 may be anything from justa few millivolts (essentially zero distance and flight time) to up totwo volts (maximum range and flight distance) depending on the distanceto the target. Time T₁₂ represents the firing of the laser as indicatedby the REF signal and T₁₃ represents the receipt of the reflected lasersignal as indicated by the RX(OUT+) signal. Transistor 210 is turned onat T₁₂ and turned off at T₁₃. As a consequence, V1 will equal V2 atanytime between T_(14A) (minimum distance when T₁₂ and T₁₃ areessentially coincident) and T_(14B) (maximum range of the laser rangefinder 10). Times T_(14A) through T_(14B) represent the range of times(depending on the distance to the target) when the value of V1 isdischarged below the level of V2 and the comparator 236 output changesstate stopping the timer.

The actual laser flight time LASER_(TIME) (or FLIGHT_(TIME)) then equalsT_(14A) (or T_(14B)) minus T₁₀ minus ZERO_(TIME) or, T₁₄ minus T₁₃. Thetime T₈ has to be greater than T₃, and T₁₄ is greater than or equal toT₃. There is no theoretical limit on the lower range of the laser rangefinder 10 and flight time (and distance) can be measured down to zerodue to its linearity. The only factors in the near zero range are thetime it takes transistor 210 to turn on, the propagation time of thelaser beam and the various circuit gates, but since the time for each ofthese factors is the same during calibration as during flight time, theyessentially cancel out. The precision timing section 34 can beeffectively utilized down to on the order of ten nanoseconds and stillremain perfectly linear. RANGE to a target is then a constant, “k” timesthe quantity FLIGHT_(TIME)−ZERO_(TIME) over CAL_(TIME)−ZERO_(TIME).

For each of the values: ZERO_(TIME), CAL_(TIME) and FLIGHT_(TIME) valuesare accumulated and are expressed in time units that derive from thevery accurate crystal oscillator 30. Typically, ten pulses may beutilized to establish the ZERO_(TIME) average, ten pulses to establishthe CAL_(TIME) average and ten pulses to establish the minimum precision(or rough) FLIGHT_(TIME) range to the target. Another group of tenthrough thirty laser pulse FLIGHT_(TIME)s may be also averaged in orderto obtain a higher precision distance to a target as indicated by a“precision flag” which may be displayed on the LCD display 32 within thelaser range finder 10 eyepiece. Nevertheless, the actual values derivedin these time expansions will, of course, vary with time, temperatureand aging and affects the gain of the transistors, the leakages, as wellas the value of the resistances and capacitances. Initially the exactvalues of these effects are completely unknown but, through the use ofthe zero and calibration functions above-described, the zero problem hasbeen eliminated, and a crystal reference calibration has been providedfor the entire flight time without having to resort to a complicatedcounter circuitry.

Another aspect of the precision timing section 34 is the automatic setnoise control and invertor 168 provides, in conjunction with othercircuit elements, a hardware hold off function. Upon firing of the laserand receipt of the reference signal REF on line 94 at the CLK input offlip-flop 158, a certain time must elapse, as determined by the timeconstant of resistor 164 and capacitor 166, before the D input goeshigh. Until that time, all noise pulses and/or early laser pulses on theclock line are ignored. The purpose for this function is that, when thelaser fires, it generates unintended ground bounce and noise that mayprematurely rigger the receive flip-flop 162 rather than the real laserreturn signal (RX(OUT+). For that reason, a hold off period is providedcorresponding to the minimum range of the laser range finder 10 and, asan example, considering a minimum range of about twenty yards, theholdoff time is approximately 60 nanoseconds. With a lower sensitivitylaser range finder 10 utilized at shorter ranges the function can beeliminated and it is clearly most useful with a high sensitivityreceiver where the noise from the firing circuit determines an effectiveminimum range.

Transistor 174 provides an additional function and allows themicrocomputer 270 to extend the hold off range by asserting the HOLD OFFsignal on line 258. In this manner, the minimum range of the laser rangefinder 10 may be extended out to, for example, sixty or eighty yards,whatever is the desirable setting. This microcomputer 270 hold offfunction may be implemented by the mode switch 126 and would allowshooting through branches, twigs, precipitation or other partialobstructions. Be extending the hold off range out beyond such partialobstructions, there is insufficient back scatter from the obstructionsto trigger the precision timing section 34 and the measurement will bemade to the desired target instead of the intervening obstructions. Thisis accomplished by now allowing flip-flop 162 to trigger until a settimer period has elapsed. Transistor 174 is the switching deviceutilized to allow setting of an extension to the hold off range and gate180 is used to determine the receive pulse width in conjunction with thedischarge rate of capacitor 194. This allows the microcomputer 270,which has a built in analog-to-digital (“A/D”) convertor, to determinethe residual voltage on capacitor 194 and therefore derive a measure ofthe pulse width, (which is a measure of the return signal power) andthus use an internal lookup table to correct for that power variationand get a higher range accuracy. When the logic reset signal /RESET online 156 is low, transistor 190 clamps capacitor 194 to the +5 voltrail. During the laser measurement routine, the transistor 190 is turnedoff. When a pulse subsequently arrives, that bit turns on transistor 200and the voltage in capacitor 194 will be discharged via resistor 198 forthe duration of that pulse. The charge on capacitor 194 is thendigitized by the processor to determine the effect of incoming power.

With reference additionally now to FIG. 8, the automatic noise thresholdsection 36 of the laser range finder 10 is shown. The automatic noisethreshold section 36 receives the RX(OUT+) signal from the laser receivesection 22 (shown in FIG. 1) on line 100 for input thereto throughresistor 314. Resistor 314 is connected to the anode of diode 316 whichhas its cathode connected to the “+” input of operational amplifier(“OpAmp”) 318 forming a V₃ node 320. V₃ node 320 is coupled to circuitground through the parallel combination of resistor 322 and capacitor324. The output of OpAmp 318 is coupled back to the “−” input thereof aswell as to line 102 through resistor 326 for supplying the V_(threshold)signal to the laser receive section 22 (shown in FIG. 1). Line 102 isconnected through resistor 330 to the center tap of potentiometer 332which has one terminal thereof connected to a source of +5 volts throughresistor 334 and another terminal thereof coupled to circuit groundthrough resistor 336.

Lines 308 and 310 from the microcomputer 270 (shown in FIG. 6) areconnected through resistors 338 and 330 respectively to line 102.Additionally, line 312 from microcomputer 270 is connected to line 102through resistor 342 as shown.

In operation, the automatic noise threshold section 36 in conjunctionwith the CPU section 28 (shown in FIG. 6) provides a simply implementedyet highly effective threshold adjustment to the laser receive section22 (shown in FIG. 3) As shown in FIG. 3, the laser receiving diode 24utilizes a high-voltage source (of about 50 volts) supplied via a noisefiltering network, comprising low pass filter network 106, to bias it.The diode 24 responds with an output current proportional to theincoming laser light which is generally a short duration laser pulseproducing a short current pulse which is amplified by transistors 118,120, 122, 124, comprising the active circuit elements of atransimpedance amplifier 116. The transimpedance amplifier 116 producesan output voltage pulse proportional to the incoming laser pulseimpinging on the laser receiving diode 24. The output of thetransimpedance amplifier 116 is capacitively coupled to the “+” input ofcomparator 134, which is a high speed comparator. When the laser pulseinput to the “+” input crosses a threshold determined by the voltage onthe “−” threshold pin, a positive output pulse is produced.

To maximize performance, the threshold of the comparator 134 has to beset for maximum sensitivity in order detect the weakest possible laserpulse to get the maximum performance out of the laser range finder 10.Conventional approaches include using digital controls or apotentiometer to adjust the threshold. However, these approaches havethe down side that over time and temperature changes the gain of thereceiver will change with the background noise generated by thebackground light rendering a fixed threshold as less than an idealsolution.

The automatic noise threshold section 36 of FIG. 8 discloses a circuitthat automatically sets a threshold such that a constant noise pulsefiring rate is output from the detector comprising resistor 314, diode316, capacitor 324 and resistor 322. In operation, when the thresholdpin of the comparator 134 (FIG. 3) is at a considerably higher voltagethan the input pin, no noise pulses will appear at the output due to theinherent amplifier and optically generated noise. As the voltages on thethreshold and input pins are brought closer together, noise pulses willappear at the output and, when the voltage levels are nearly coincident,a great deal of noise can be seen. In essence then, the automatic noisethreshold section 36 sets the noise pulse rate at that point at which,given the right firmware algorithm, one can still acquire the target andnot be blinded by the noise. The higher the noise that can be tolerated,and the closer the voltage levels at the threshold and input pins of thecomparator 134, the weaker the laser pulse that can be detected. Theautomatic noise threshold section 36 automatically adjusts thatthreshold level to maintain constant noise pulse firing rate.

As shown in FIG. 8, this is accomplished by monitoring the digital logicreceive signal RX(OUT+) on line 100 that goes to the receive flip-flop162 (shown in FIG. 4). The detector monitors line 100 for the presenceof noise pulses via a detector comprising the aforementioned resistor314, diode 316, capacitor 324 and resistor 322. The value of resistor322 is typically considerably greater than that of 314, on the order ofa 150:1 ratio. The peak amplitude of the noise pulses is typically at ornear the logic threshold, except for very narrow pulses where thecomparator will not reach full amplitude, however, the width of thesepulses is going to vary randomly because it depends on the noise signalthat is being detected. Moreover, the spacing of the noise pulses willalso vary at a random rate, but, for any given threshold setting, therewill be a fixed average rate. The average rate is dependent on thethreshold. Therefore, during the time the pulse is high, capacitor 324charges via resistor 314 and diode 316 at a rate determined by the highon the logic pulse, resistor 314 and whatever voltage is still existingor capacitor 324.

Initially, capacitor 324 is charged as follows. once the noise pulseterminates, the logic line goes back to zero. There is a residualvoltage on capacitor 324, diode 316 will be reverse biased, and thedischarge path is now via resistor 322. (As previously described, thevalue for resistor 322 is chose to provide a relatively longer timeconstant, a factor of 150.) When another pulse comes in, capacitor 324will charge a bit more. What will then happen is, quite rapidly, (i.e.within a few milliseconds) the voltage across capacitor 324 stabilizesat a rate that is proportional to the average firing rate. The reasonfor having a large ratio between resistor 314 and resistor 322 isbecause the noise pulses typically may average 50 nanoseconds wide, andthe averaged time between them to maximize the sensitivity of the laserrange finder 10 should be of the order of the two microseconds or so. Asan example, of a 50% voltage were desired, and the high state wasoccurring for 50 nanoseconds while the low state average was occurringfor one microsecond, a 20:1 ratio would be produced. Nevertheless, theoptimum ratio has been determined empirically to be about 150:1 aspreviously described and is related to average pulse widths (typicallyon the order of 30 nanoseconds in length) and pulse repetition rates (onthe order of 4 microseconds) with a typical voltage level of 1.5 volts.

Op amp 318 is configured as a unity gain buffer, although it need not beunity gain, with a voltage V3 at its “+” input pin on node 320. Theinput is high impedance and the output is low impedance in order todrive external circuitry. The voltage that is derived at the output ofthe op amp 318 is then fed into a resistor network comprising resistor338, resistor 340, resistor 342 and resistor 330. A summing node of theresistor network on line 102 goes to the threshold control to providethe signal V_(threshold) to the laser receive section 22 (shown in FIG.3). Resistor 330 is connected to the center tap of a potentiometer 332so that the DC voltage on the other end of resistor 330 can becontrolled.

In combination, the circuit comprises a feedback network such that, ifthere are no noise pulses, then V3 is zero and V_(threshold) and dropsto a low value. Initially, V_(threshold) will be higher, and the “−”input of comparator 134 (shown in FIG. 3) will be higher than the “+”input, forcing a logic low on the output as the starting state. As thelevel of V3 on node 320 falls, the voltage level on the “−” pin ofcomparator 134 starts approaching the level of the signal from thetransimpedance amplifier 116 on the positive “+”. When it approaches thenoise zone, noise pulses start appearing. As soon as noise pulses startappearing, a charge appears on node 320, so V3 stops to charge up, andwhen the two match, that's the feedback point and it stops. Basically,the voltage on the threshold is set at such a point that the noisefiring rate maintains V3 at that voltage which is necessary to maintainV_(threshold). Because very small changes in V_(threshold) make a verylarge change in the noise firing rate, typically, a ten millivolt changein V_(threshold) will change the voltage V3 at node 320 by about a volt.What is produced then, is a fairly high gain feedback loop, such thatV_(threshold) will track very closely the noise firing rate and V3 willstabilize very accurately and rapidly. This further provides thecapability to adjust the noise firing rate by controlling the bias andforcing V3 to compensate. The voltage V3 at node 320 then represents thenoise firing rate.

NSET1 line 308 and NSET2 line 310, are two control lines from themicrocomputer 28 such that when held low or high, adjusts the noise rateto obtain the maximum range to different reflectivity targets. If bothlines 308 and 310 are taken high, V3 will drop to compensate to maintaina constant threshold noise. Similarly, potentiometer 332 provides anadjustment such that the threshold point may be set together with thelevel of V3. Typically, the V3 point might be set equal to: 0.5, 1.0,1.5 and 2.0 volts as desirable choices for the average noise firingrates. As such, since resistor 338 is approximately twice the value ofresistor 340, four voltage combinations are obtained roughly equallyspaced in voltage by half a volt. Potentiometer 332 is used to set thefirst voltage level to 0.5 or the last one to 2.0 while the intervalsare determined by the logic control lines 308 and 310 set NSET1 andNSET2. Obviously, this approach could be extended, four combinationsprovides adequate resolution in the particular implementation of thelaser range finder 10 described and shown. When both lines 308 and 310are high, there is a current injected into the node comprising theV_(threshold) line 102, and to compensate for that, V3 must drop, soless current flows through resistor 326 and vice versa. V3 will followthese values, depending on the permutations of logic high and lowsignals on the lines 308 and 310. Resistor 330 is used just to set wherethis whole block resides while potentiometer 332 is used to establishthe initial set point. Since the noise characteristics from unit to unitwill vary somewhat, potentiometer 332 enables the setting of the initialdevice characteristics.

Resistor 342 is of a considerably lower value than resistors 338 and 340and its value is chosen such that, when the REFLECTOR MODE signal online 312 is asserted by being taken high, V3 will drop to zero and willstay there because it cannot go below zero. At this point, the feedbackloop is saturated and is no longer effective, so V_(threshold) is nolonger stabilized. In operation, line 312 will be pulled high by aconsiderable voltage, on the order of 0.4 volts, such that it completelydesensitizes the laser receive section 22 so the laser range finder 10will then only respond to a retro reflector. In this mode of operationthe receiver is detuned and its noncooperative range drops from 500yards down to about 30 or 40 yards, such that the laser range finder 10only latches onto a retro reflector or survey prism comprising a highgrade reflector that returns the laser energy back to the source.Possible applications also include determining the distance to aparticular golf hole where a laser reflector is attached to the pin andthe signal might otherwise be actually returned from trees behind or infront of the green in a more sensitive mode of operation.

The essence of the automatic noise threshold section 36 is, aspreviously described, a feedback loop comprising the detected averagenoise firing rate forming a feedback loop that controls the threshold.Use of this circuit has resulted in an addition of almost 50% to therange of the laser range finder 10 compared to attempting to manuallyset the threshold. By setting the noise firing rate, noise pulses arebeing produced deliberately, all the time and the only way you to takeadvantage of that fact is by implementing a firmware algorithm in themicrocomputer 270 that allows you to discriminate between noise pulsesand laser return pulses. What the algorithm does is, during the laserfiring process, on the first pulse that fires, it gets a laser pulse,and it places it in a stack of pulses. For example, the stack may havelocations designated 0 through 9, to enable 10 pulses to be maintainedin the stack. The values of the FLIGHT_(TIME) are saved, corrected forpower return, (the microcomputer 270 determines the power level of thereturn signal and corrects the flight time for power return) and placedin one of the locations in the stack. Upon receipt of the next pulse,the microcomputer 270 will then compare the next pulse with theremaining locations in the stack. Initially, most of the locations willbe empty, and there will be no match. If no match is found, themicrocomputer 270 puts the pulse in the stack and carries on, merelyplacing pulses in the stack, and then when it gets to the top, it goesback and overwrites the base, so you have a history of N number ofpulses in the stack. Any time a new pulse comes in, it compares theentire stack for a match, where N=10, it searches the preceding tenpulses for a match.

The reason for doing that is, since a high noise firing rate has beendeliberately set to get maximum sensitivity, many noise pulses are goingto have shown up, but the noise pulses will be of random occurrence andthe chance of a precision match is very low. Because the tolerance canbe set as any other firmware parameter, a default value will betypically loaded that has been determined empirically. As an example, atolerance of a few nanoseconds may be set for a match to be assumed tobe a real target and not a noise pulse. Utilizing the algorithm, theprocess continues, trying to lock on the target until a match isachieved. The match need only be two pulses within the preset tolerance(providing very acceptable results) or, if higher sensitivity weredesired, a match of three through N may be specified, depending on thereliability needed to guarantee a real target and not a noise pulse. Inan exemplary operation, the first pulse (pulse 0) could be the realtarget, followed by eight noise pulses, and as long as the ninth pulseis again the real target, the distance to the target can be accuratelydetermined. The stack can be increased in size up to whatever memorylimit is available in the system, depending on how far into the noiselevel the laser range finger 10 must work.

Having found a match, the average of the match values may then be usedto compare all subsequent pulses, rather than needing to place them in astack and only pulses that match up with that initial match average willcontribute to the measurement. If a certain number of pulses elapsebefore another matching pulse is received, it may be assumed that anaccidental lock-on to noise has been achieved and the process restarts.By adjusting the various parameters, a trade off can be made between thetime it takes to get a measurement to how far into the noise the laserrange finger 10 must work. Because the noise rate can set to whatever isdesired by means of the automatic noise threshold section 36, it ispossible to optimize the algorithm to provide the optimum acquisitioncharacteristics against time and against range.

The higher the value of V3, the more noise is coming out of thereceiver, and the more sensitive the laser receive section 22 isrunning. The probability of a noise pulse showing up is proportional tothe flight time, so given a very “black” target, the maximum range willbe less, but the maximum flight time is also less, so a higher noiserate can be tolerated. Therefore, running at a higher gain will providethe best range to a black target. On the other hand, if the target veryreflective, a high gain is not required, so the noise rate can belowered, which then provides the same probability of a noise pulseappearing over a longer flight range, and therefore a quick acquisitionon a bright white target can be achieved. Thus, by depressing the modeswitch 126, different modes of operation of the laser range finder 10can be selected. As an example, one mode might be utilized to find therange to reflective road signs out to a distance of 1000 yards or more.Alternatively, aiming the laser range finder 10 at something like wetblack tree bark, might reduce the maximum range to only 350-400 yardsand so a different operational mode might be selected which wouldotherwise require a relatively long time to hit the road sign, if ever,because there would always be a noise pulse in the way. The mode switchallows the setting of these variables to maximize the range of the laserrange finder 10, depending on the target quality and a visual indicationof the target quality selected may be provided to the operator on theinsight, LCD display 32 wherein the first mode would correspond to thebrightest target or most reflective target, and the Nth mode wouldcorrespond to the least reflective target.

While there have been described above the principles of the invention inconjunction with specific apparatus, it is to be clearly understood thatthe foregoing description is made only by way of example and not as alimitation on the scope of the invention.

What is claimed is:
 1. A laser range finder comprising: a lasertransmitting section for producing a series of transmitted laser pulsesdirectable towards a target and producing a plurality of returned laserpulses at least partially reflected therefrom in response thereto; alaser receiving section for receiving said plurality of returned laserpulses and noise pulses, said laser receiving section comprising a lasersignal receiving device coupled to an input of a transimpedanceamplifier, said transimpedance amplifier providing an amplified outputsignal of said laser signal receiving device for input to a comparatorcircuit for providing an automatic noise threshold adjustment to saidlaser receiving section to facilitate discrimination between saidreturned laser pulses and said noise pulses; a central processingsection coupled to said laser transmitting and laser receiving sectionsfor determining a distance to said target based on a time of flight ofsaid transmitted and returned laser pulses; and a user viewable displaycoupled to said central processing section for displaying said distanceto said target.
 2. The laser range finder of claim 1 wherein said lasersignal receiving device comprises a laser receiving diode.
 3. The laserrange finder of claim 1 wherein said comparator circuit comprises aprecision comparator.
 4. The laser range finder of claim 3 wherein saidprecision comparator comprises a MAX 913 precision comparator.
 5. Thelaser range finder of claim 1 wherein said transimpedance amplifier iscapacitively coupled to an input of said comparator circuit.
 6. Thelaser range finder of claim 5 wherein said input of said comparatorcircuit is further coupled to a voltage divider network comprising firstand second resistors coupling a supply voltage source to a referencevoltage level.
 7. The laser range finder of claim 1 wherein saidcomparator circuit sets a substantially constant noise threshold firingrate.
 8. The laser range finder of claim 7 wherein said substantiallyconstant noise threshold firing rate is set by means of aresistor/capacitor network.
 9. The laser range finder of claim 8 whereinsaid resistor/capacitor network couples an output of said comparatorcircuit to a threshold voltage input thereof.
 10. The laser range finderof claim 9 wherein said resistor/capacitor network comprises at leastone resistive element coupled to said output of said comparator circuitand coupled in series with at least one capacitive element coupled to areference voltage level, said resistive and capacitive elements having acircuit node therebetween.
 11. The laser range finder of claim 10wherein said circuit node is coupled to said threshold voltage input ofsaid comparator circuit.
 12. The laser range finder of claim 11 whereinsaid laser signal receiving device is further coupled to a power supplysource through a low pass filter network.
 13. The laser range finder ofclaim 12 wherein said low pass filter network comprises at least oneresistive element coupling said power supply source to a first terminalof said laser signal receiving device, a second terminal of said lasersignal receiving device being coupled to said input of saidtransimpedance amplifier, said low pass filter network also comprisingat least one capacitive element coupling said first terminal of saidlaser signal receiving device to a reference voltage level.